Semiconductor integrated circuits with power reduction mechanism

ABSTRACT

This invention is to reduce the power dissipation of a semiconductor integrated circuit chip when it is operated at an operating voltage of 2.5 V or below. To achieve the object, a switching element is provided in each circuit block within the semiconductor integrated circuit chip. The constants of the switching element are set so that the leak current in the switching element of each circuit block in their off-state is smaller than the subthreshold current of the MOS transistors within the corresponding circuit block. The active current is supplied to the active circuit blocks, while the switching elements of the non-active circuit blocks are turned off. Thus, the dissipation currents of the non-active circuit blocks are limited to the leak current value of the corresponding switching elements. As a result, the sum of the dissipation currents of the non-active circuit blocks is made smaller than the active current in the active circuit blocks. Since the dissipation currents of the non-active circuit blocks can be reduced while the active current is caused to flow in the active circuit blocks, the power dissipation in the semiconductor integrated circuit chip can be reduced even in the active state.

This is a continuation of application Ser. No. 08/620,686, now U.S. Pat.No. 5,606,065 filed Mar. 21, 1996; which is a continuation ofapplication Ser. No. 374,990 now U.S. Pat. No. 5,521,527 filed Jan. 19,1995; which is a continuation of application Ser. No. 178,020 filed Jan.6, 1994 (now U.S. Pat. No. 5,408,144).

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is relevant to U.S. patent application Ser. No.07/972,545 filed on Nov. 6, 1992 in the names of T. Kawahara et al., nowU.S. Pat. No. 3,224,601 and U.S. patent application Ser. No. 08/045,792filed on Apr. 14, 1993 in the names of M. Horiguchi et al., abandonedthe contents of which are incorporated herein by reference.

BACKGROUND OF THE INVENTION AND SUMMARY OF THE INVENTION

The present invention relates to semiconductor integrated circuitssuitable for high-speed and low-power operation, and particularly to asemiconductor integrated circuit formed of small-geometry MOStransistors.

The semiconductor integrated circuits have so far been developed towardthe scaling down of MOS transistors. However, since the minute structureof MOS transistors reduces their breakdown voltage the more as thedegree of the minuteness becomes greater, the operating voltage of thesmall-geometry MOS transistors must be lowered, as described inInternational Symposium on VLSI Technology, Systems and Applications,Proceedings of Technical Papers, pp. 188-192 (May 1989). The operatingvoltages of the semiconductors used in the battery-operated portableelectronic apparatus must be further reduced for their low powerconsumption.

In order to maintain their high-speed operation under reduced operatingvoltages, it is also necessary to decrease the threshold voltage (V_(T))of the MOS transistors. The reason for this is that the operating speedis governed by the effective gate voltage of the MOS transistors, or theremainder of the subtraction of V_(T) from the operating voltage, orthat it increased with the increase of this effective gate voltage. Forexample, in a 16-gigabit DRAM which is expected to have 0.15 μm or belowin effective channel length, about 4 nm in gate oxide film thickness, 1V in standard operating voltage within chip and about 1.75 V in boostedword line voltage, the constant current threshold voltage of transistorsis calculated to be -0.04 V. The term, constant current thresholdvoltage of transistors is the gate-source voltage under the conditionsof a ratio, 30 of effective channel width to effective channel lengthand a drain current of 10 nA. In this case, the substrate-source voltageis 0, the junction temperature is 25° and a typical condition isassumed. For simplicity, the threshold voltage of p-channel MOStransistors is shown with the opposite sign.

When V_(T) is reduced, however, the drain current cannot be completelycut off due to the drain current characteristic of the subthresholdregion of MOS transistors. This problem will be described with referenceto FIG. 22A which shows a conventional CMOS inverter. When the inputsignal IN to the CMOS inverter has a low level (=V_(ss)), the n-channelMOS transistor M_(N) is turned off. When the input signal IN has a highlevel (=V_(cc)), the p-channel MOS transistor M_(P) is turned off.Therefore, in either case, from the ideal point of view, no currentflows from the high source voltage V_(cc) through the CMOS inverter tothe low source voltage V_(ss), or ground potential.

When the threshold voltage V_(T) of the MOS transistors is reduced,however, the subthreshold characteristic cannot be neglected. As shownin FIG. 22B, the drain current I_(DS) in the subthreshold region isproportional to the exponential function of the gate-source voltageV_(GS), and expressed by the following equation (1). ##EQU1## where W isthe channel width of the MOS transistors, I₀ and W₀ are the currentvalue and channel width used when V_(T) is defined, and S is thesubthreshold swing (the reciprocal of the gradient of the V_(GS) -logI_(DS) characteristic). Thus, the drain current in the subthresholdregion (, or the subthreshold current) flows even under V_(GS) =0. Thesubthreshold current can be expressed by the following equation (2).##EQU2##

When the input signal to the CMOS inverter shown in FIG. 22A is notchanged, or when it is not operated, the off-state transistor of theCMOS inverter is at V_(GS) =0. Therefore, the current flowing from thehigh source voltage V_(cc) through the CMOS inverter to the low sourcevoltage V_(ss), or ground potential is the current I_(L) which flows inthe off-state transistor of the CMOS inverter.

This subthreshold current, as shown in FIG. 22B, is exponentiallyincreased from I_(L) to I_(L) ' when the threshold voltage is decreasedfrom V_(T) to V_(T) '.

Although the increase of V_(T) or the reduction of S reduce thesubthreshold current as will be seen from the equation (2), the increaseof V_(T), incurs the reduction of the speed due to the decrease of theeffective gate voltage, while the reduction of S will be difficult forthe following reason.

The subthreshold swing S can be expressed by using the capacitanceC_(OX) of the gate dielectric and the capacitance C_(D) of the depletionregion under the gate as in the following equation (3). ##EQU3## where kis the Boltzmann constant, T is the absolute temperature and q is theelementary charge. As will be seen from the equation (3), the conditionof S≧kTl_(n) 10/q is limited for any values of C_(OX) and C_(D). Thus,it is difficult for S to be reduced to 60 mV or below at roomtemperature (about 300 k).

Thus, in the semiconductor integrated circuit including MOS transistorswith a low value of V_(T) the amount of DC current consumption ofnon-operating circuits is remarkably increased because of the phenomenonmentioned above when the operating voltage becomes low (for example, 2 Vor 2.5 V). Particularly, upon high-temperature operation, S becomeslarge, making this problem further serious. In the downsizing age offuture computers and so on, when reduction of power is important, theincrease of the subthreshold current becomes a substantial problem.

This problem will be further considered taking a memory, which is atypical semiconductor integrated circuit, as an example. The memorygenerally includes, as shown in FIG. 23, a memory array MA, an X decoder(XDEC) and word driver (WD) for selecting and driving a row line (wordline W) for the selection of a memory cell MC within the memory arrayMA, a sense amplifier (SA) for amplifying the signal on a column line(data line D), a sense amplifier driving circuit (SAD) for driving thesense amplifier, a Y decoder (YDEC) for selecting a column line, and aperipheral circuit (PR) for controlling these circuits. The main partsof these circuits are designed based on the CMOS inverter logic circuitmentioned above.

When the threshold voltage V_(T) of transistors (hereinafter, forsimplicity the absolute values of the threshold voltages of thep-channel and n-channel MOS transistors are assumed to be equal toV_(T)) is low, a subthreshold current, that is a current flowing in thesource-drain path of the MOS transistors of which the V_(GS), issubstantially 0.

Therefore, the sum of the subthreshold currents becomes particularlygreat in the circuits having a large number of MOS transistors, such asdecoders, drivers or the peripheral circuit section.

For example, in the decoders or drivers, a small number of particularcircuits are selected from a large number of circuits of the same typeby the address signal, and driven. FIG. 24 shows an example of theconventional word driver for DRAM.

If the threshold value V_(T) of the MOS transistors of all CMOS drivers#1-#r is large enough, the subthreshold current, that is a currentflowing in the source-drain paths of the MOS transistors ofsubstantially zero V_(GS), is substantially zero in each of a largenumber of nonselected circuits. In general, the number of the decoderand driver is increased with the increase of the storage capacity of thememory. However, even though the storage capacity is increased, thetotal current is not increased unless the subthreshold current flows inthe circuits that are not selected in the decoders or drivers.

If the threshold voltage V_(T) is decreased as mentioned above, however,the subthreshold current increases in proportion to the number ofnonselected circuits.

In the prior art, when the chip is in the standby mode (nonselectedstate), almost all the circuits within the chip are turned off so thatthe power dissipation can be reduced as much as possible. However, it isnot possible any more to reduce the power current dissipation even inthe standby mode because the subthreshold current flows when the MOStransistors are highly scalled down.

When V_(T) is small, the subthreshold current that is a current flowingin the source-drain paths of the MOS transistors with V_(GS) beingsubstantially zero, causes a trouble not only in the standby mode butalso in the operating mode. Generally the current I_(ACT) flowing whenthe chip is in the active mode and the current I_(STB) flowing when thechip is in the standby mode are respectively expressed by

I_(ACT) ≈I_(OP) +I_(DC), and

I_(STB) ≈I_(DC),

where I_(OP) is the charging and discharging current to and from theload capacitance of the circuits within the chip, and given by

    I.sub.OP =C.sub.TOT ·V.sub.cc ·f

in which V_(cc) is the operating voltage of the chip, C_(TOT) is thetotal load capacitance of the circuits within the chip, and f is theoperating frequency. In addition, I_(DC) is the subthreshold currentgiven above. The subthreshold current is exponentially increased withthe decrease of V_(T) as indicated by the equation (2).

So far, since V_(cc) is large and V_(T) is also large enough, thecondition of

    I.sub.OP >I.sub.DC

can be satisfied. Therefore, the following equations can be given:

    I.sub.ACT ≈I.sub.OP,

    I.sub.STB ≈I.sub.DC

In this case, I_(DC) is substantially zero. Thus, for I_(ACT), only theincrease of I_(OP) has been considered to cause a problem.

However, since I_(DC) is increased with the decrease of V_(cc) andV_(T), finally the following condition is satisfied:

    I.sub.OP ≦I.sub.DC

In addition, it is found that if V_(cc) and V_(T) are decreased, thefollowing condition is given:

    I.sub.OP <I.sub.DC

In this case, the expressions of

    I.sub.ACT ≈I.sub.DC,

    I.sub.STB ≈I.sub.DC

can be given. Therefore, the increase of the subthreshold current I_(DC)also becomes a problem to the current I_(ACT) which flows when the chipis operating.

FIG. 25 shows an example of the results predicted for the currentdissipation in the DRAM. This prediction is made at a junctiontemperature of 75° C. with typical conditions. From FIG. 25, it will beseen that the I_(DC) of the 4-G bit DRAM exceeds I_(OP) when itsoperating voltage is assumed to be 1.2 V.

If we considers the worst conditions, the subthreshold current I_(DC)causes a problem even when the effective channel length, gate oxide filmthickness and operating voltage are respectively about 0.25 μm, 6 nm and2.5 V. Here, the values corresponding to a 256-M bit DRAM are used. Inthe prior art, when the operating voltage is 3.3 V, the constant currentthreshold voltage defined as the gate-source voltage of the transistorof which the ratio of the effective channel width to effective channellength is 30 and in which the drain current is 10 nA is one tenth of theoperating voltage, or 0.33 V at a substrate-source voltage of 0 volt anda junction temperature of 25° C. with typical conditions. At this time,the extrapolated threshold voltage defined as the gate-source voltagewhen the drain current characteristic of a saturated region isextrapolated for zero current is about 0.2 V higher than the constantcurrent threshold voltage, or about 0.53 V. When the operating voltageis reduced to 2.5 V, the extrapolated threshold voltage is reduced toabout 0.4 V in proportion to the operating voltage in order to assurethe effective gate voltage. Since the difference between theextrapolated threshold voltage and the constant current thresholdvoltage is substantially constant, the constant current thresholdvoltage is about 0.2 V. In addition, the temperature dependency of thethreshold voltage must be considered. In general, when the operation ofthe chip at room temperature is assured, it must be guaranteed at anormal ambient temperature T_(a) of 0° C. through 70° C. Moreover, thejunction temperature T_(j) within the chip can be found from theequation of

    T.sub.j =T.sub.a +θ.sub.ja ·P.sub.d

where P_(d) is the dissipation power, and θ_(ja) is the thermalconductivity of the chip, and thus a higher temperature must beconsidered. If the source voltage and the active current I_(ACT) are 2.5V and 50 mA, respectively, and if T_(a) is 75° C. including a marginwhen θ_(ja) is 200° C./W, the junction temperature T_(j) is 100° C. Theconstant current threshold voltage at this value of T_(j) is about 0.1 Vwhen the temperature dependency of the threshold voltage is assumed tobe -1.6 mV/°C. In addition, if we consider the threshold voltagevariation due to the process dispersion as 0.1 V, the constant currentthreshold voltage with 10 nA at the worst condition is about 0.0 V. Inthis case, if the effective gate length is 0.25 μm, the gate width whichis used when defining the constant current threshold voltage is about7.5 μm. If the total value of gate widths of the MOS transistors withinthe chip which contributes to I_(DC) is about 4 m, the subthresholdcurrent I_(DC) is found to be 5 mA from the equation (2) when thesubthreshold swing S is 100 mV/dec. This value of 5 mA corresponds toone tenth of I_(ACT) as assumed above, and is thus too large to beneglected. Therefore, when the operating voltage is about 2.5 V orbelow, the subthreshold current in the CMOS logic circuit causes aproblem.

We now consider the dissipation power of a CMOS LSI which is demandedfor its application. The dissipation power in the LSI used in portableapparatus should be treated as the average dissipation power in theperiod of time in which it is energized and thus it includes both thestandby current and the active current. Particularly in thebattery-operated apparatus, both the currents are important because theidle time is determined by the average dissipation power. There are anumber of LSIs which keep operating almost within the period of time inwhich they are energized, such as IC processors, memories and ASICs forhigh-speed operation. In these LSIs, only the active current isimportant, and thus the condition of I_(ACT) ≈I_(STB) may be allowed. Ineither case, the reduction of I_(ACT) is an important subject in allLSIs. In the prior art, since I_(DC) is dominant for I_(ACT) in the nearfuture as described above, it becomes important to reduce the leakage(penetrating) current flowing through the CMOS LSI in the active mode.

Accordingly, it is an object of the invention to provide a semiconductorintegrated circuit capable of reducing the dissipation power of thesemiconductor integrated circuit including circuits which haveenhancement type MOS transistors which operate at an operating voltageof 2.5 V or below and which cause a significant current in thesource-drain path when V_(GS) is substantially zero.

It is another object of the invention to reduce the subthreshold currentof the MOS transistors included in the word driver, decoder, senseamplifier driving circuit and so on of a memory or a semiconductorintegrated circuit in which the memory is incorporated.

In order to achieve the above objects, when some circuits of thesemiconductor integrated circuit are operated to change their outputs,the subthreshold current of the MOS transistors in the other circuitswhich are not changed in their outputs is reduced so that thedissipation power can be decreased.

More specifically, according to this invention, there is provided asemiconductor integrated circuit chip operating at an operating voltageof 2.5 V or below, comprising:

a first terminal at which a first operating potential is applied;

a second terminal at which a second operating potential is applied;

a first circuit block coupled between the first terminal and the secondterminal; and

a second circuit block coupled between the first terminal and the secondterminal;

wherein the first circuit block permits an active current to flowbetween the first terminal and the output terminal when the firstcircuit block responds to an input signal to its input terminal toproduce an output signal at its output terminal,

wherein the second circuit block includes a plurality of subcircuitblocks each of which includes a MOS transistor having its sourceconnected to a first node and its gate connected to the input terminaland a load having one end connected to the drain of the MOS transistorand the other end connected to a second node,

wherein the MOS transistor of each of the plurality of subcircuit blockscauses a subthreshold current in its source-drain path when thegate-source voltage is substantially 0,

wherein the plurality of first nodes of the plurality of subcircuitblocks are coupled through a plurality of switching elements to thefirst terminal, and the plurality of second nodes of the plurality ofsubcircuit blocks are coupled to the second terminal,

wherein constants of said plurality of switching elements are set sothat the leak currents of the plurality of switching elements in theiroff-state are smaller than the subthreshold current of the MOStransistor of the corresponding one of said plurality of subcircuitblocks, and

wherein the current dissipation in each of the plurality of subcircuitblocks of the second circuit block is limited to a leak current value ofthe corresponding one of the plurality of switching elements by turningoff the plurality of switching elements so that the sum of the currentdissipations in the plurality of subcircuit blocks is made smaller thanthe active current of the first circuit block.

Therefore, even when the semiconductor integrated circuit chip isactive, the first circuit block operates within the chip, while thesubthreshold current in the non-active second circuit block can bereduced.

In addition, according to this invention, there is provided asemiconductor integrated circuit chip operating at an operating voltageof 2.5 V or below, comprising:

a first terminal at which a first operating potential is applied;

a second terminal at which a second operating potential is applied;

a first circuit block coupled between the first terminal and the secondterminal; and

a second circuit block coupled between the first terminal and the secondterminal;

wherein the first circuit block permits an active current to flowbetween the first terminal and the output terminal when the firstcircuit responds to an input signal to its input terminal to produce anoutput signal at its output terminal,

wherein the second circuit block includes a plurality of subcircuitblocks each of which includes a MOS transistor having its sourceconnected to a first node and its gate connected to the input terminaland a load having one end connected to the drain of the MOS transistorand the other end connected to a second node,

wherein the MOS transistor of each of the plurality of subcircuit blockscauses a subthreshold current to flow in its source-drain path when thegate-source voltage is substantially 0,

wherein the plurality of first anodes of the plurality of subcircuitblocks are coupled through a plurality of switching elements to thefirst terminal, and the plurality of second nodes of the plurality ofsubcircuit blocks are coupled to the second terminal,

wherein constants of said plurality of switching elements are set sothat the leak currents of said plurality of switching elements in theiroff-state are smaller than the subthreshold current of the MOStransistor of the corresponding one of said plurality of subcircuitblocks, and

wherein the current dissipation in each of the plurality of subcircuitblocks of the second circuit block is limited to a leak current value ofthe corresponding one of the plurality of switching elements by turningoff the plurality of switching elements so that the sum of the currentdissipations in the plurality of subcircuit blocks is made smaller thanthe active current of the first circuit block,

wherein each of the plurality of subcircuit blocks of the second circuitblock causes an active current to flow between the first terminal andthe output terminal of each of the subcircuit blocks when eachsubcircuit responds to an input signal to its input terminal to producean output signal at its output,

wherein the first circuit block includes a plurality of subcircuitblocks, each of which includes a MOS transistor having its sourceconnected to the first anode and its gate connected to the inputterminal, and a load having its one end connected to the drain of theMOS transistor and the other end connected to the second anode,

wherein the MOS transistor of each of the plurality of subcircuit blocksof the first circuit block causes the subthreshold current to flow inits source-drain path when the gate-source voltage is substantially 0,

wherein the plurality of first anodes of the plurality of subcircuitblocks of the first circuit block are coupled through the plurality ofswitching elements to the first terminal, and the plurality of secondanodes of the plurality of subcircuit blocks of the first circuit blockare coupled to the second terminal,

wherein the constants of the plurality of switching elements of thefirst circuit block are fixed so that the leak currents of the pluralityof switching elements of the first circuit block in their off-state aresmaller than the threshold current of the MOS transistor of thecorresponding one of the plurality of subcircuit blocks of the firstcircuit block, and

wherein the plurality of subcircuit blocks of the second circuit blockare made active by turning on the plurality of switching elements of thesecond circuit block, and the current dissipation in each of theplurality of subcircuit blocks of the first circuit block is limited toa leak current value of the corresponding one of the plurality ofswitching elements by turning off the plurality of switching elements ofthe first circuit block so that the sum of the current dissipations inthe plurality of subcircuit blocks is made smaller than that of theactive currents of the second circuit block.

Therefore, when the semiconductor integrated circuit chip is active, thefirst circuit block within the chip is operated, while the subthresholdcurrent in the second non-active circuit block can be reduced. Inaddition, while the second circuit block is operated within the chip,the subthreshold current in the first non-active circuit block can bereduced.

Moreover, according to this invention, there is provided a semiconductorintegrated circuit chip comprising:

a plurality of first circuit blocks;

a plurality of first switching elements;

a first operation potential power line coupled common to the pluralityof first switching elements; and

a second switching element coupled between the first operation potentialpower line and a first operation potential point,

wherein each of a plurality of first nodes of the plurality of firstcircuit block is coupled to the first operation potential power linethrough the corresponding one of the plurality of first switchingelements,

wherein a plurality of second nodes of the plurality of first circuitblocks are coupled to a second operation potential power line,

wherein each of the plurality of first circuit blocks includes a MOStransistor having its source connected to the corresponding one of thefirst nodes and its gate connected to an input terminal, and a loadhaving its one end connected to an drain of the MOS transistor and theother end connected to the corresponding one of the second nodes,

wherein the MOS transistor of each of the plurality of first circuitblocks causes a subthreshold current to flow in its source-drain pathwhen the gate-source voltage is substantially 0,

wherein constants of the plurality of first switching elements are setso that the leak current of each of the plurality of first switchingelements in its off-state is smaller than the subthreshold current ofthe MOS transistor of the corresponding one of the plurality of firstcircuit blocks,

wherein the current dissipation of each of the plurality of firstcircuit blocks is limited to a leak current value of a corresponding oneof the plurality of first switching elements by turning off theplurality of first switching elements, and

wherein a constant of the second switching element is set so that theleak current of the second switching element in its off-state is smallerthan the sum of the leak currents of the plurality of first switchingelements, and thus the sum of current dissipations in the plurality offirst circuit blocks is limited to the leak current value of the secondswitching element.

Therefore, the current dissipation of each of the plurality of firstcircuit blocks in the standby mode is limited to the subthresholdcurrent of the first switching elements or below. In addition, the sumof the current dissipations in the plurality of first circuit blockscoupled to the first operation potential line in the standby mode islimited to the subthreshold current of the second switching element orbelow.

The total sum of the sum of the dissipation currents in the plurality offirst circuit blocks and the sum of the dissipation currents in theplurality of second circuit blocks is limited to the subthresholdcurrent of the fifth switching element or below.

The concept common to the semiconductor integrated circuit chips givenin the summary of the invention is to have a plurality of circuit blocksand at least two circuit terminals through which a desired operatingvoltage is applied to these circuit blocks, and to have current controlmeans provided for the subthreshold currents in the circuit blocksbetween each of the circuit blocks and at least one of the circuitterminals, so that the leakage current in a certain one of the circuitblocks is controlled by the current control means during a periodincluding part of the time in which at least another one of the circuitblocks logically operates. Thus, when the semiconductor integratedcircuit chip itself is in the operating state, during the period inwhich a certain one of the circuit blocks is operated, the leakagecurrent in another non-operated one of the circuit blocks can bereduced. As a result, it is possible to reduce the total leakage currentof the semiconductor integrated circuit chip in the active mode.

Therefore, even if the threshold voltage of the MOS transistor isdecreased as it is scaled down, the leakage current flowing in thenonselected circuit can be minimized.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing an arrangement for limiting the time inwhich the subthreshold current flows.

FIG. 2 is a timing chart for controlling the arrangement shown in FIG.1.

FIG. 3 shows an embodiment of a one-dimensional arrangement of blocks.

FIG. 4 shows an embodiment of a two-dimensional arrangement of blocks.

FIG. 5A shows an embodiment of a one-dimensional arrangement of worddriver blocks.

FIG. 5B is a timing chart for the operation of the embodiment shown inFIG. 5A.

FIG. 6 is a graph showing the operating point of the p-channel MOStransistor of the word driver in the embodiment shown in FIG. 5A.

FIG. 7 is a graph showing the block-number dependency of the leakagecurrent for one-dimensional selection.

FIG. 8 shows an embodiment of a one-dimensional arrangement of senseamplifier driving circuit blocks.

FIG. 9 shows an example of the arrangement of a main part of the memoryarray.

FIG. 10 shows the effect of the invention.

FIG. 11 shows an embodiment of a one-dimensional arrangement of decoderblocks.

FIG. 12 shows another embodiment of a one-dimensional arrangement ofword driver blocks.

FIG. 13 shows an embodiment of a one-dimensional arrangement ofn-channel MOS driver blocks.

FIG. 14 shows an embodiment of a two-dimensional arrangement of atypical selection system.

FIG. 15 is a timing chart for controlling the embodiment shown in FIG.19.

FIG. 16A shows an embodiment of a two-dimensional arrangement of worddriver blocks.

FIG. 16B is a timing chart for the operation of the embodiment shown inFIG. 16.

FIG. 17 is a graph showing the operating point of the p-channel MOStransistor of the word driver in the embodiment shown in FIG. 16.

FIG. 18A is a graph showing the block-number dependency of the leakagecurrent for two-dimensional selection with the sector number being usedas a parameter.

FIG. 18B is a graph showing the block-number dependency of the leakagecurrent for two-dimensional selection with the circuit number within ablock being used as a parameter.

FIG. 19 shows an arrangement of the division of 512 word drivers intofour blocks.

FIG. 20 shows an embodiment of a two-dimensional arrangement of decoderblocks.

FIG. 21 shows an embodiment of the application to the hierarchical typeword line arrangement.

FIG. 22A is a circuit diagram of the conventional CMOS inverter.

FIG. 22B is a graph showing the subthreshold characteristic of thetransistor.

FIG. 23 is a block diagram of a memory.

FIG. 24 shows a conventional method of supplying current to worddrivers.

FIG. 25 is a graph showing the prediction of current dissipation in DRAMby the prior art.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

First, a description will be made of an application of the invention toa word driver (WD shown in FIG. 23) of DRAM. We consider the state aftera word line has been selected and supplied with a necessary word voltageV_(CH). In the conventional arrangement shown in FIG. 24, if V_(T) isonly high enough, almost no subthreshold current flows in thesource-drain paths of the MOS transistors of all CMOS drivers of whichthe gate-source voltages are substantially zero. However, when V_(T) isreduced to about 0.4 V or below, the subthreshold current flows in theword drivers. As, the capacity becomes large and the word driver number(r) increases, the intensity of this current becomes significant. Thetotal subthreshold current I_(A) can be expressed as ##EQU4## whereV_(T) is the threshold voltage defined by the current value I₀, and S isthe subthreshold swing as shown in FIG. 22B. The word driver sourceV_(CH) is normally produced by boosting the external source voltagewithin the chip, and thus its current driving ability is limited.Therefore, the increase of I_(A) cannot be treated.

There are three methods for the counter-measure: (1) a necessary voltageis applied to the power line to the word driver for a desired period oftime, (2) the word driver group is divided into a large number of blockseach of which is formed of a plurality of drivers, and a necessaryvoltage is applied only to a particular block which is desired toselect, and (3) both methods given above are combined.

FIG. 1 shows an example of the method in which a necessary voltage isapplied to the power line to a word driver for a desired period of time,thereby limiting the time in which the subthreshold current flows. Thismethod has the feature that a desired word voltage is applied to acommon power line of a block after the logical input to the driver hasbeen fixed. The operation of the PMOS transistor constituting the worddriver will be mentioned with reference to the timing chart of FIG. 2.In the known memory cells for DRAM which are formed of NMOS transistorsand capacitors, the voltage of all nonselected word lines must be V_(ss)(0 V), and hence the PMOS transistors within all word drivers includingword drivers to be selected have the gate voltage V_(CH). Then, whenselecting operation begins, only the gate N_(x1) of the PMOS transistorof the selected driver #1 becomes 0 V. At this time, the other drivers#2 through #r remain at V_(CH). Thus, the gate voltage is fixed for thePMOS transistors of all the word drivers. Before the gate voltage of thePMOS transistors is fixed, the voltage on the common power line P_(B)which is connected with the sources of the PMOS transistors is set at acertain lower voltage than V_(CH), or in an extreme case, at 0 V inorder that the subthreshold current in the PMOS transistors can beneglected. Here, the certain voltage given above is so selected as to beabout V_(CH) -(0.4 V-V_(T)) as compared with V_(T) for the PMOStransistor. This is because the effective gate voltage resulting fromthe subtraction of V_(T) from the gate-source voltage is, as describeabove, necessary to be about 0.4 V in order to make the subthresholdcurrent of the PMOS transistor negligibly small. For example, in a16-gigabit DRAM, since V_(CH) =1.75 V and V_(T) =0.04 V as describedabove, here the certain voltage is about 1.31 V. When the common powerline P_(B) is increased to V_(CH) after the gate voltage is fixed, thevoltage V_(CH) is applied from the corresponding PMOS transistor to theselected word line. When the gate voltages of the PMOS transistors inall the word drivers are set to V_(CH) after the voltage application fora desired period of time, the selected word line discharges to 0 Vthrough the corresponding PMOS transistor. Then, the voltage on thecommon power line P_(B) is again reduced to the certain voltage or belowas mentioned above. In the period of time in which the common drivingline is separated from V_(CH) by this driving method, there is nosubthreshold current. However, during the time in which the voltageV_(CH) is applied to the common driving line, the subthreshold currentstill flows in the PMOS transistors of the nonselected word drivers.When any word driver is operated, V_(CH) must be applied to the commondriving line, and hence the subthreshold current cannot be reduced.

Even when the logic input to the driver is fixed after a necessary wordvoltage is applied to the common power line, a correct voltage can beproduced on the word line. In this case, during the period of time inwhich the logic input to the driver is fixed after the word voltage isapplied to the power line, the subthreshold current wastefully flows inall the word drivers. On the other hand, in the method of applying aword voltage to the common power line after the logic input is fixed,the waste current during this period can be reduced. However, theoperation speed is lowered. Since the parasitic capacitance of thecommon power line is large, the rise time on this line becomes long andthus the access time increases the more.

FIG. 3 and FIG. 4 show conceptional embodiments for solving the aboveproblems. These embodiments have the feature that the word driver groupis divided into a large number of blocks each of which is formed of aplurality of drivers, and that the subthreshold current is caused toflow only in a selected block. In other words, the subthreshold currentupon operation can be reduced in inverse proportion to the block number.FIG. 3 shows a one-dimensional arrangement of m blocks of n word driverseach (where m·n=r). In this embodiment, the subthreshold current can bereduced by m-fold that in the embodiment shown in FIG. 1. FIG. 4 shows atwo-dimensional arrangement (matrix) of k blocks (k is not Boltzmannconstant) of l word drivers each in the row direction and j blocks of lword drivers each in the column direction (where j·k·l=r). In thisarrangement, the subthreshold current can be reduced by 1/(j·k) that inthe embodiment shown in FIG. 1. Some embodiments concerning theone-dimensional arrangement and two-dimensional arrangement will bedescribed in detail.

A specific embodiment of the one-dimensional arrangement will be firstdescribed in detail. FIG. 5A shows an embodiment of the one-dimensionalarrangement of word driver blocks, and FIG. 5B is a timing chart for theoperation of this embodiment. In this embodiment, a second power line isprovided common to the power lines to the blocks so as to behierarchical. This hierarchical type power line system has the followingtwo features.

(1) Hierarchical type power lines to driver blocks: m blocks of n worddrivers #1 through #n each are provided, power lines P₁ through P_(m) tothe blocks are respectively connected through block selectiontransistors Q₁ through Q_(m) to a common power line P, and the line P isconnected to a power line of word voltage V_(CH) through a transistor Qfor selecting one of the operating mode and the standby mode.

(2) Setting of hierarchical gate width: the gate width (a·W) of theblock selection transistors Q₁ through Q_(m) is previously selected tobe much smaller than the sum (n·w) of gate widths of the word drivertransistors within one block (a<<n), and the gate width (b·W) of thetransistor Q is previously selected to be much smaller than the sum(m·a·W) of the gate widths of the selected transistors of all blocks(b<<m·a).

Upon operation, the transistors Q and Q1 are turned on, permitting thevoltage V_(CH) to be applied to the power line P1 for the block B1including the selected word driver #1. Here, the voltages V_(T) of allthe transistors are assumed to be equal and a low value. According tothis arrangement, the subthreshold current of each of the nonselectedblocks B₂ through B_(m) is smaller than that of the corresponding blockselection transistor Q₂ through Q_(m) because the subthreshold currentis proportional to the gate width of the transistor. Even if a currentof n·i tends to flow, all the subthreshold currents are limited by thesubthreshold current (a·i) of the block selection transistor. At thattime, the voltages on the power lines P₂ through P_(m) to thenonselected blocks still remain reduced by Δ V substantially as in thestandby mode because the subthreshold currents of the transistors Q₂through Q_(m) for charging the lines P₂ through P_(m) are relativelysmall. Therefore, the total subthreshold current I_(A) is calculatedfrom Table 1 to be substantially (n+m·a)i when m >>1. If a is selectedto be about 4, it is possible to reduce the effect on the operatingspeed of the series transistors Q, Q₁ and the chip area.

                  TABLE 1                                                         ______________________________________                                                Operating                                                                             Standby                                                               current current   Necessary                                                                              charge                                             (I.sub.A)                                                                             (I.sub.S) t.sub.1  t.sub.2                                    ______________________________________                                        Prior Art m · n · i                                                             m · n · i                                                             0      0                                        This      n · i + (m -                                                                   b · i                                                                          C · ΔV'                                                               C.sub.1 · ΔV              invention 1) · a · i                                        ______________________________________                                    

If the word driver number (m·n) is constant, the subthreshold currentn·i of the selected block is inversely proportional to the totalsubthreshold current m·a·i of the nonselected blocks. Thus, in order toreduce I_(A), it is desired that the subthreshold current n·i of theselected block be substantially equal to the total subthreshold currentm·a·i of the nonselected blocks, or n≈m·a. If this condition is notsatisfied, I_(A) will be increased by the amount corresponding to thedeviation from the condition. The number, m of blocks to be fixeddepends on the decoder for the block selection signal and the layoutarea. It is desirable to make I_(A) less than about five times theminimum value. To meet this requirement, a larger one of thesubthreshold current of the selected block and the total subthresholdcurrent of the nonselected blocks should be within 10 times the valuewhich minimizes I_(A). In other words, it is necessary that the ratiobetween the subthreshold current of the selected block and the totalsubthreshold current of the nonselected blocks be within the range from0.01 to 100.

FIG. 7 shows the active-mode subthreshold current I_(A) with respect tothe block number when the word driver number (m·n) is 128K. In thisgraph, the interconnection resistance and interconnection capacitanceare neglected. The calculated curve and the simulated curve exhibit thesame tendency and have the minimum value when m and n are selected tosatisfy n≈m·a. Taking a=b=4 as an example, the leakage current becomesthe minimum, or less than 1/85 the value in the prior art when the blocknumber m is selected to be 128 or 256. The tendency that the simulatedcurve takes lower values than the calculated value will be probablycaused by the effect of the on-resistance of the PMOS transistor used asa switch.

In the standby mode, all the transistors Q, Q₁ through Q_(m) are almostturned off. The total subthreshold current I_(s) becomes equal to thesubthreshold current of the transistor Q, and can be reduced by a/m·n ascompared with the prior art. The voltage on the power line to the blocksis reduced from V_(CH) by Δ V which is determined by the ratio betweenm·n·W and b·W and the subthreshold swing. At this time, the operatingpoint of the p-channel MOS transistor of the word driver is as shown inFIG. 6.

When the transistor Q is not used, or when the switch shown in FIG. 3 isreplaced by the transistor of gate width a·W, the total subthresholdcurrent m·a·i flows in m PMOS transistors of gate width a·W in thestandby mode. Since the gate width ratio a cannot be reduced even underthe divided blocks because of the driving ability of the selectingcircuit in the active state, the standby current is increased with theincrease of the block number m. When the word driver number (m·n) isconstant, n cannot be decreased unless m is increased, and hence the n·ipart of the subthreshold current in the operating mode cannot bedecreased. Therefore, the subthreshold currents in both the standby modeand the operating mode cannot be simultaneously minimized.

In this embodiment, this problem is solved by inserting the transistor Qand setting the gate width at b<<m·a so as to make a double-stagecurrent limiting function. In other words, a transistor of gate widthb·W is provided common to m transistors of gate width a·W so that thecurrent limiting function has two stages. Thus, the subthreshold currentin the operating mode remains n·i+(m-1)·a·i, while the subthresholdcurrent in the standby mode can be reduced to b·i. Since thesubthreshold current in the standby mode is not dependent on m and n, mand n can be combined to minimize the subthreshold current in theoperation mode, or n≈m·a can be established.

The operation will be described with reference to the timing chart ofFIG. 5B. In the standby mode (Φ, Φ₁ through Φ_(m) : V_(CH)), thetransistors Q and Q₁ through Q_(m) are almost turned off. Therefore, thepower line P is at a lower voltage V_(CH) -ΔV' than V_(CH), and thepower lines P₁ through P_(m) are at a further lower voltage V_(CH) -ΔVthan that. All the word lines are fixed at V_(ss) independently of thevoltage of the power lines P₁ through P_(m). When the external clocksignal/RAS (where "/" indicates a complementary signal) is turned on,the transistor Q is turned on at Φ, permitting the parasitic capacitanceC of the line P to be charged for time t₁, to reach V_(CH). Then, thetransistor Q₁ is turned on at Φ₁, permitting the parasitic capacitanceC₁ of the line P₁ to be charged for time t₂ to reach V_(CH). At thistime, the transistors Q₂ through Q_(m) are still substantially in theoff-state. Then, the word driver #1 is selected by the X decoder outputsignal X₁, and drives the word line. When the clock signal /RAS isturned off, the transistors Q and Q₁ are turned off. After lapse of along time, the power lines P and P₁ are respectively reduced to V_(CH)-ΔV' and V_(CH) -ΔV by the mechanism mentioned above. Here, the powerlines (P, P₁) can be charged to V_(CH) without reducing the access timebecause ΔV' is as small as several hundred m V even if the parasiticcapacitance C is large and because the charging time t₁ for the line Pcan be maintained to be long enough immediately after the clock signal/RAS is turned on. In addition, because of the block arrangement, theparasitic capacitance C₁ is relatively small and thus the charging timet₂ for the line P₁ can be reduced.

If the hierarchical type power lines are provided for the decoders, thesubthreshold current can be greatly reduced.

FIG. 8 shows a hierarchical power line scheme applied to a senseamplifier driving circuit (SAD in FIG. 23), and FIG. 9 shows a main partof a memory array of memory cells each of which is formed of a singletransistor and a single capacitor. Since the well-known V_(cc) /2precharge scheme is used, this sense amplifier driving circuit operatesbased on the central value, V_(cc) /2. Thus, it has the feature that thehierarchical type power lines are used for both V_(cc) and V_(ss). Here,it is assumed that the conductance of the PMOS transistor Q_(P) is equalto that of the NMOS transistor Q_(N). The CMOS sense amplifier (SA)group within the sub-array is selectively driven by the correspondingsense amplifier driving circuit. At this time, the current I_(A) 'flowing through the power lines V_(cc), V_(ss) is governed by thesubthreshold currents of a large number of nonselected driving circuits.Even if the gates of the transistors Q_(P), Q_(N) shown in FIG. 8 arerespectively applied with V_(cc) and 0 not to be selected, the senseamplifier driving lines CP, CN are at V_(cc) /2, and thus thesubthreshold current flows from P'₁ to P"₁. In order to prevent this, itis absolutely necessary to apply the hierarchical power-line scheme toboth sides. If the hierarchical type power lines are applied to onlyV_(cc), the subthreshold current of the transistor Q_(N) flows fromV_(cc) /2 to P"₁, thus reducing the level of V_(cc) /2 because thecurrent driving ability of the V_(cc) /2 supply circuit built in thechip is small.

FIG. 10 shows the active current reducing effect of the inventionapplied to the word drivers, decoders and sense amplifier drivingcircuit with the subthreshold current being assumed not to flow in theperipheral circuit (PR in FIG. 23) portion. A 16 gigabit DRAM is givenas an example. The parameters used are as follows. The threshold voltageV_(T) which is defined by a voltage at which a current of 10 nA flows ina gate width 5 μm is -0.12 V, the subthreshold swing S is 97 mV/dec, thejunction temperature T is 75° C., the effective gate length L_(eff) is0.15 μm, and the gate oxide film thickness T_(OX) is 4 nm. In addition,the word voltage V_(CH) is 1.75 V, the source voltage V_(cc) is 1 V, thecycling time is 180 ns, the refresh cycle number is 128k, the chip sizeis 23 mm×45 mm, and the total capacitance of data lines which charge anddischarge in one cycle is 17 nF. According to this invention, the activecurrent can be reduced from about 1.05 A to about 1/10 that value in theprior art, or to 109 mA. This is because the subthreshold current can begreatly reduced from about 0.97 A to about 1/30 that value in the priorart, or to 34 mA.

The embodiments of the invention have been described in which the worddrivers and sense amplifier driving circuit are provided in theone-dimensional arrangement of blocks. This invention is not limited tothe above embodiments, but may take various modifications as givenbelow.

FIG. 11 shows an example of the hierarchical type power-line schemeapplied to the decoder. This arrangement shown in FIG. 1 has ANDcircuits each of which is formed of two stages, or a NAND circuit and aninverter of a CMOS logic circuit. As illustrated in FIG. 11, the featureis that the hierarchical power lines are provided on both sides ofV_(cc) and V_(ss), though the operating point is not V_(cc) /2 unlikethe sense amplifier driving circuit. In the standby mode, all the NANDcircuits produce V_(cc), and in the operating mode, a small number ofones of the NAND circuits produce 0 V. Since the subthreshold current isdetermined by the n-channel MOS transistor on V_(ss) side, thehierarchical power lines are on the V_(ss) side. On the contrary, allthe inverters produce 0 V in the standby mode, and a small number ofones of the inverters produce V_(cc) in the operating mode. Since thesubthreshold current is determined by the p-channel MOS transistor, thehierarchical power lines are used on the V_(cc) side.

This invention can be applied to the circuit group which produces thesame voltage in the standby mode, and a small number of circuits ofwhich operate in the operating mode. At this time, all the circuits areof the same transistor size, but may be different in their construction.

FIG. 12 shows another embodiment of the application of the invention tothe word driver. In the embodiment shown in FIG. 12, 16 ones of 2 M worddrivers are simultaneously operated. This embodiment is different fromthe embodiment shown in FIG. 5A in that a plurality of power lines areused. As illustrated in FIG. 12, each block is formed of 512 worddrivers, and 8 sectors S₁ through S₈ are provided each of which isformed of 512 blocks (B₁,1 through B₁,256, B₂,1 through B₂,256). Twoblocks (for example, B₁,1 and B₂,1) of each sector share one power line(for example, P₁). The first 128 ones of the power lines P₁ through P₂₅₆are connected through block selection transistors Q₁ through Q₁₂₈ to apower line P_(L), and the other 128 ones are connected through blockselection transistors Q₁₂₉ through Q₂₅₆ to a power line P_(R). Thesepower lines P_(L) and P_(R) are common to the 8 sectors. The power linesP_(L) and P_(R) are further connected through transistors Q_(L) andQ_(R) to the power line V_(CH). The gate width of the transistors Q₁through Q₂₅₆ is selected to be much smaller than the total gate width ofthe transistors of the word drivers of two blocks, or 1×10³ worddrivers. In addition, the gate width of transistors Q_(L), Q_(R) isselected to be much smaller than the total gate width of the blockselection transistors connected to each of the power lines P_(L), P_(R),or 8×128 block selection transistors. In the operating mode, the eightsectors make the same operation. For example, the transistors Q_(L),Q_(R) and the transistor Q₁ within each sector are turned on and V_(CH)is applied to the two blocks (B₁,1 and B₂,1) including the selected worddriver #1. The subthreshold current is the same as when m and n arerespectively selected to be 256 and 4×10³ in the embodiment shown inFIG. 5. Thus, when a plurality of circuits are simultaneously operated,a plurality of blocks are selected at a time. If the transistors asswitches are divided into a plurality of groups and connected, the powerlines can be shortened so that the interconnection resistance can bereduced. Thus, the power line P₁ to the selected block can be charged ina short time.

FIG. 13 shows an example of the application of the invention ton-channel MOS drivers. This embodiment has the feature that thehierarchical type power lines are connected to the drains of thetransistors. Each driver is a push-pull circuit configuration of twon-channel MOS transistors. The nonselected drivers produce 0 V, and theselected driver produces V_(cc) -V_(T). The hierarchical type powerlines are provided on the drain, or V_(cc) side of the transistors.Thus, the subthreshold current can be reduced without changing theoutputs of the nonselected drivers as in the embodiment shown in FIG. 5.In FIG. 13, if the block selection transistors Q₂ through Q_(m) are inthe off state, the voltages on the power lines P₂ through P_(m) aregreatly reduced so that no current flows in the word driver transistorseven though the subthreshold current is not so affected by the drainvoltage. Thus, this invention can be applied to the logic circuit otherthan the CMOS one.

In either of the above embodiments, it is desired that the substrate ofthe transistors which has not been mentioned be connected to the powersupply. In this case, the amount of charge necessary for charging thepower lines becomes smaller and the charging time is shorter, than whenthe substrate is connected to the power lines which are connected to thedrains. For example, in the embodiment shown in FIG. 5, all thesubstrate of the p-channel MOS transistors is connected to V_(CH), sothat when the voltage V_(CH) of the power lines of the nonselectedblocks is reduced by ΔV as described above, the threshold voltage of thep-channel MOS transistors within the nonselected blocks is increased bythe body effect. The source voltage becomes lower than the gate voltage,and in addition since the threshold voltage is increased, the samecurrent reducing effect can be achieved by a small ΔV as compared withthe case in which the substrate is at the same voltage as the drain.

In the above embodiments, the threshold voltages of the transistors areall equal. If the threshold voltage of the transistors used as switchesis made higher than the other transistors, the subthreshold current canbe further reduced. If the threshold voltage of the transistors Q and Q₁through Q_(m) shown in FIG. 5 is made higher than that of thetransistors within the word drivers and if a and b are selected to belarge, the operating speed can be prevented from being reduced by theon-resistance of the switches, and the subthreshold current can befurther reduced. The subthreshold current is exponentially affected bythe threshold voltage of the switches at the off-state, while theon-resistance is only linearly affected. Even if the gate capacitance isincreased with the gate width, there is no problem in the operatingspeed provided that the charging time t₁, t₂ is assured in FIG. 7. Inaddition, as to the layout area, the number is relatively small and thusthere is no problem. In some case, even if only the transistor Q has ahigh threshold voltage, the standby current can be effectively reduced.

As shown in the timing chart of FIG. 7, during the active period inwhich the clock signal /RAS is 0 V, the transistors Q and Q₁ are kept onwith Φ and Φ₁ reduced. This is realized by controlling the Φ by a signalfor specifying the operating mode at the active time and standby timewhich is produced by the clock signal /RAS and by controlling the Φ₁ bythe combination of that signal and an address signal. In addition, byuse of a signal for specifying the period from the trailing edge of theclock /RAS to the end of the word line driving, it is possible to makethe Φ and Φ₁ V_(CH) and transistors Q and Q₁ off after the word linedriving. Thus, the subthreshold current after the word line driving canbe reduced to the same extent as the standby current I_(s) even at theactive time. This effect is the greater, the longer the active period inwhich the clock signal /RAS is 0 V. In this case, for rewriting thememory cell, it is necessary to reduce the Φ and Φ₁ and turn on thetransistors Q and Q₁ for a constant period from the leading edge of theclock signal /RAS. Even in the embodiment in which this invention isapplied to the decoders shown in for example FIG. 11, the subthresholdcurrent after fixing the output can be further reduced.

Some embodiments of this invention concerning the two-dimensionalarrangement will be described in detail. FIG. 14 shows an embodiment ofa typical selection system of the two-dimensional arrangement. FIG. 15is a timing chart for the operation of this embodiment. A necessary wordvoltage V_(CH) is applied to a row line (P_(s1)) corresponding to, forexample, a block B₁,1 which is desired to select, and 0 V is applied tothe corresponding column (Φ_(B1)). The block selection p-channel MOStransistor Q₁,1 is turned on, and the power line P₁,1 belonging to B₁,1is charged to V_(CH). Since the p-channel MOS transistor constitutingthe word driver belonging to B₁,1 already has a fixed gate voltage,V_(CH) is applied to the word line selected according to this. Ofcourse, as described above, even if the gate voltage is fixed after theapplication of V_(CH) to P₁,1, the word line can be correctly driven.After the voltage is applied for a desired period of time, the powerline P₁,1 is discharged to 0 V through the n-channel MOS transistorconnected thereto. The power line belonging to the nonselected blocksremains 0 V. Here, for simplicity, we consider that the V_(T) of theblock selection p-channel MOS transistor and power line dischargingn-channel MOS transistor is selected to be high enough (about 0.4 V).Since the power line for the nonselected blocks is always 0 V, nosubthreshold current flows in the word drivers of the nonselectedblocks. Therefore, the total subthreshold current can be greatly reducedto only the subthreshold current of the selected block. In addition,since the power line is divided into power lines with a small-parasiticcapacitance which are driven, higher speed operation than in theembodiment shown in FIG. 1 can be realized.

FIG. 16A is another embodiment of the selection system of thetwo-dimensional arrangement, and FIG. 16B is a timing chart for theoperation of this embodiment. In this embodiment, a plurality of thearrangement for the one-dimensional selection shown in FIG. 5A areprovided, and transistors are further provided between the power supplyand the power terminals. In this case, the transistor Q in FIG. 5corresponds to the transistors Q_(s1) through Q_(sj) in FIG. 16A. In thesame manner as shown in the embodiment of FIG. 14, only the block at anintersection is selected by a row power line (for example, P_(s1)) and acolumn control line (for example, Φ_(B1)). This embodiment is differentfrom the embodiment shown in FIG. 14 in the point given below. In FIG.14, when each block is not selected, the power line of each block is at0 V, and even when the block selecting operation is started, all thepower lines for the nonselected blocks are at 0 V. When either one ofthe blocks is selected, that power line must be charged from 0 V toV_(CH), and thus there is the drawback that the operation speed is lowand that the transient current becomes large. In order to solve thisproblem, when a certain block is shifted from the nonselected state tothe selected state, it is desired that the voltage change of that powerline be suppressed to be as small as possible and that the subthresholdcurrent of the other nonselected blocks is suppressed to be negligiblysmall. The embodiment shown in FIG. 16A is able to realize thisoperation, and has two features as follows.

(1) The hierarchical type power lines to blocks of drivers each: j·kblocks of a single word driver each are provided in a matrix shape.These blocks are divided into j groups, or sectors of k blocks each. Thepower lines P_(B1) through P_(Bk) for the blocks of each sector areconnected through block selection transistors Q_(B1) through Q_(Bk) tothe power line of the sector (for example, P_(s1)). In addition, thepower line P_(s1) through P_(sj) of each sector is connected through asector selection transistor Q_(s1) through Q_(sj) to the power line P.Moreover, the power line P is connected through the transistor Q forselecting one of the operating mode and the standby mode to the powerline of word voltage V_(CH).

(2) The setting of hierarchical gate width: the gate width (d·W) of theblock selecting transistor is selected to be much smaller than the totalgate width (l·W) of the word driver transistor within the block (d <<l).In addition, the gate width (e·W) of the sector selection transistor isselected to be much smaller than the total gate width (k·d·W) of theblock selection transistor within the sector (e<<k·d). Moreover, thegate width (f·w) of the transistor Q is selected to be much smaller thanthe total gate width (j·e·W) of all sector selection transistors(f<<j·e).

Upon operation, the transistors Q, Q_(s1), Q_(B1) are turned on,permitting V_(CH) to be applied to the power lines P_(B1) and P_(S1)corresponding to the block B₁ including the selected word driver (#1)and the sector S₁ including B₁. Here, V_(T) of all transistors isassumed to be the same low value. Thus, the total subthreshold currentof the nonselected sector (S₁ through S_(j)) equals to the subthresholdcurrent of the corresponding single sector selection transistor (Q_(s2)through Q_(sj)). In addition, the subthreshold current of each of thenonselected blocks (B₂ through B_(k)) within the selected sector (S₁)equals to the subthreshold current of the corresponding single blockselection transistor (Q_(B2) through Q_(Bk)). This is because thesubthreshold current is proportional to the gate width of thetransistor. Therefore, even though a current of l·i tends to flow in thenonselected blocks within, for example, the sector S₁, the totalsubthreshold current is limited by the subthreshold current (d·i) of theblock selection transistor. As a result, the total subthreshold currentI_(A) is substantially (1+k·d+j·e) i when k>>1, j>>1 as listed on Table2. If d, e and f are selected to be about 4, the effect on the operationspeed of the series transistor (Q, Q_(s1), Q_(B1)) and the chip area canbe reduced.

When the word driver number (j·k·l) is constant, the product of thesubthreshold current l·i of the selected block within the selectedsector, the total subthreshold current k·d·i of the nonselected blockswithin the selected sector and the total subthreshold current j·e·i ofthe nonselected sectors is constant. Therefore, in order to reduceI_(A), it is necessary that the subthreshold current l·i of the selectedblock within the selected sector, the total subthreshold current k·d·iof the nonselected blocks within the selected sector and the totalsubthreshold current j·e·i of the nonselected sectors be set atsubstantially the same value, or l≈k·d≈j·e. The total subthresholdcurrent I_(A) is increased with the increase of the deviation from thiscondition. The setting of the sector number j and block number k dependson the decoders within the selected circuit and the layout area. Thetotal subthreshold current I_(A) is desired to be set at about fivetimes or below as large as the minimum value. To this end, the largestone of the subthreshold current of the selected block within theselected sector, the total subthreshold current of the nonselectedblocks within the selected sector and the total subthreshold current ofthe nonselected sectors should be selected to be within 15 times thevalue at which the total subthreshold current I_(A) is the minimum.Thus, the ratio of the largest one to the smallest one of the threesubthreshold currents, or the subthreshold current of the selected blockwithin the selected sector, the total subthreshold current of thenon-selected blocks within the selected sector and the totalsubthreshold current of the nonselected sectors should be less thanabout 60 (15÷(√15)).

FIGS. 18A and 18B show graphs of the dependency of the operatingsubthreshold current I_(A) on the sector number and block number whenthe word driver number (m·n) is 32×10⁶. In FIG. 18A, the block number kand word driver number l are changed for a particular sector number j.In FIG. 18B, the sector number j and block number k are changed for theword driver number l within a particular block. In addition, in FIG.18A, the one-dimensional selection shown in FIG. 5A is also shown forreference. The coefficients d, e and f of the gate width are selected tobe 8, and the interconnection resistance and interconnection capacitanceare neglected. When j=256, K=128 and l=1K so that l≈k·d≈j·e, the minimumcan be obtained. At this time, it can be reduced to 1/8K the value inthe prior art, or 1/8 the minimum value in the one-dimensionalselection.

In the standby mode, all the transistors Q, Q₁ through Q_(k) are madealmost in the off-state. The total subthreshold current I_(s) equals tothe subthreshold current of the transistor Q, and thus it can be reducedby f/j·k·l as compared with the prior art. The voltage on the power lineof the block is reduced from V_(CH) by ΔV which is determined by theratio between j·k·l·W and f·W and subthreshold swing as shown in FIG.17. This ΔV is different from that in FIG. 6.

The transistor Q is provided so that the subthreshold currents in boththe standby mode and operating mode can be minimized at a time like thetransistor Q in the embodiment shown in FIG. 5A. Thus, the currentlimiting function becomes a three-stage function. The subthresholdcurrent in the standby mode is not dependent on j, k and l, and thus therelation of j, k and l can be established to be l≈k·d≈j·e under theconditions that the subthreshold current in the standby mode is smalland that the subthreshold current in the operating mode is the minimum.

Table 2 lists the current values obtained for a 16 gigabit DRAM. Theparameters used are as follows. The threshold voltage V_(T) which isdefined by a voltage at which a current of 10 nA flows in a gate widthof 5 μm is -0.12 V, the subthreshold swing S is 97 mV/dec., the junctiontemperature T is 75° C., the effective gate length L_(eff) is 0.15 μm,the gate oxide film thickness T_(ox) is 4 nm, the word voltage V_(CH) is1.75 V, and the source voltage V_(cc) is 1 V. According to thisinvention, the subthreshold current in the operating mode can be reducedto about 1/350 as small as about 700 mA in the prior art, or about 2 mA,and the subthreshold current in the standby mode can be reduced to about1/33000 that in the prior art, or about 20 μA.

                  TABLE 2                                                         ______________________________________                                        Operating     standby  Amount of charge                                       current       current  to charge                                              (I.sub.A)     (I.sub.S)                                                                              t.sub.1  t.sub.2                                                                              t.sub.3                                ______________________________________                                        Prior art                                                                            j · k · l · i                                                 j · k · l · i                                               0      0      0                                           (695 mA)   (695 mA)                                                    This   l · i + (k -                                                                    f · i                                                                         C.sub.S1 · ΔV"                                                        C.sub.B1 · ΔV'                                                        C.sub.1 · ΔV          invention                                                                            1) · d · i + (j -                                                      (21.2 μA)                                                       1) · e · i                                                  (1.99 mA)                                                              ______________________________________                                    

(These values are expected ones of a 16 Gb DRAM.)

FIG. 18 shows operating waveforms. In the standby mode (Φ, Φ_(s1)through Φ_(sj), Φ_(B1) through Φ_(Bk) : V_(CH)), the transistors Q,Q_(s1) through Q_(sj) and Q_(B1) through Q_(bk) are substantially in theoff-state. Thus, the power line P is at a lower voltage V_(CH) -delta V"lower than V_(CH), the power lines P_(S1) through P_(Sj) at a furtherlower voltage V_(CH) -ΔV' than that, and P_(B1) through P_(Bk) at afurther much lower voltage V_(CH) -ΔV. All the word lines are fixed to avoltage V_(ss) independently of the voltage of the power lines P_(B1)through P_(Bk). When the external clock signal /RAS (here "/" indicatesthe bar signal) is turned on, the transistor Q is turned on at Φ,permitting the parasitic capacitance C of the power line P to be chargedfor time t1 to reach V_(CH). Then, the transistor Q_(s1) is turned on atΦ_(s1), permitting the parasitic capacitance C_(s1) of the power lineP_(s1) to be charged for time t₂ to reach V_(CH). In addition, thetransistor Q_(B1) is turned on at Φ_(B1), permitting the parasiticcapacitance C_(B1) of the power line P_(B1) to be charged for time t₃ toreach B_(CH). At this time, the transistors Q_(s2) through Qsj andQ_(B2) through Q_(Bk) remain almost in the off-state. Thereafter, theword driver #1 is selected by the X decoder output signal X₁, drivingthe word line. When the external clock signal /RAS is turned off, thetransistors Q, Q_(s1) and Q_(B1) are turned off. The power lines P,P_(s1) and P_(B1), after lapse of a long time, become at V_(CH) -ΔV",V_(CH) -ΔV' and V_(CH) -ΔV. Here, the power lines (P, P₁) can be chargedto V_(CH) without reducing the access time because delta V" is as smallas several hundred mv even if C is large and because the charging timet₁ for P can be maintained to be long enough immediately after theexternal clock signal /RAS is turned on. Moreover, since the arrangementis formed of sectors and blocks, the parasitic capacitance C_(s1),C_(B1) are relatively small and thus the charging time (t₂, t₃) forP_(s1), P_(B1) can be decreased.

In this embodiment, it is also desired that all substrate for thep-channel MOS transistors be connected to V_(CH) as is similar to theone-dimensional arrangement. In this case, the amount of chargenecessary for charging the power lines is smaller and the charging timecan be more reduced, than when the substrate is connected to the powerlines to which the drains are connected. As described above, when thevoltages on the power lines for the nonselected blocks are reduced by ΔVfrom V_(CH), the threshold voltages of the p-channel MOS transistorswithin the nonselected blocks are increased by the body effect. Sincethe threshold voltages are increased in addition to the fact that thesource voltage is lower than the gate voltage, the same current reducingeffect can be achieved by small ΔV as compared with the case in whichthe substrate voltage is equal to the drain voltage.

Since the word voltage V_(CH) is produced by boosting the source voltageV_(cc), a higher voltage than in the other circuits is applied to thegates of the MOS transistors of the word drivers. Thus, the V_(T) can beincreased the more and the current can be decreased. However, theoperating speed can be slightly decreased.

This drawback can be negligibly obviated by decreasing the thresholdvoltage of the transistors within the word driver and by increasing thethreshold voltage of the transistors used as switching transistors to ahigher voltage than that. If the threshold voltages of the transistorsQ, Q_(s1) through Q_(sj) and Q_(B1) through Q_(Bk) shown in FIG. 16 areincreased to be higher than those of the transistors within the worddriver and if d, e and f are set at large values, the operating speedcan be prevented from being decreased due to the on-resistance of theswitch, and also the subthreshold current can be further decreased. Thisis because the subthreshold current is exponentially affected by thethreshold voltage at the off-state, but the on-resistance is affectedonly linearly. If the charging time t₁, t₂, t₃ in FIG. 18 can be assuredirrespective of whether the gate capacitance increases with the increaseof gate width, the operating speed cannot be reduced. Therefore, thesubthreshold current can be further reduced without decreasing theoperating speed. There is also no problem with the layout area becausethe number of elements is relatively small. In some case, the standbycurrent can be reduced by using a high threshold voltage transistor forthe transistor Q.

In this embodiment, while a single p-channel MOS transistor is used as aswitch, other various elements or circuits can be considered providedthat they meet the following two conditions.

(1) When the switch is selected: when the switch is assumed to beshort-circuited, the current driving ability of the switch is greaterthan the active current (the subthreshold current and the chargingcurrent of the word line selected) flowing in the load (for example, asingle word driver for the block selecting switch) of the switch.

(2) When the switch is not selected: when the switch is assumed to beshort-circuited, the current supplying ability of the switch is smallerthan the standby current (the subthreshold current) flowing in the load.

The impedance should be changed to a low value and a large value uponselection and upon non-selection, respectively so that these twoconditions can be met.

In the operation shown in FIGS. 18A and 18B, the voltages Φ, Φ_(s1),Φ_(B1) are left low and the transistors Q, Q_(s1), Q_(B1) are kept onduring the active period in which the clock signal /RAS is 0 V. Thisoperation can be achieved if the voltage Φ is controlled by the signalwhich is produced by the clock /RAS and which specifies the operatingmode at the active time and at the standby time, and if the voltagesΦ_(s1) and Φ_(B1) are controlled by a combination signal of that signaland an address signal. In addition, by using the signal for specifyingthe period from when the clock /RAS falls off to when the driving of theword line ends, after the word line driving it is possible that thevoltages Φ, Φ_(s1), Φ_(B1) are increased to V_(CH) and that thetransistors Q, Q_(s1), Q_(B1) are turned off. Thus, even at the activetime the subthreshold current after the word line driving can be reducedto the same extent as the standby current I_(s). This effect is thelarger, the longer the active period in which the clock /RAS is 0 V. Inthis case, in order to make rewriting in the memory cell, it isnecessary that the voltages Φ, Φ_(s1), Φ_(B1) be reduced for a constantperiod of time from the leading edge of the clock /RAS so that thetransistors Q, Q_(s1), Q_(B1) are turned on.

FIG. 19 shows an arrangement in which 512 word drivers are divided intofour blocks. That is, 512 memory cells (MC₁ through MC₅₁₂) are providedfor a pair of data lines, and selected by 512 word lines. The memorycells are arranged at a high density by reducing the word line width andthe spacing between the word lines to the same extent as the featuresize. Therefore, the word drivers cannot be laid out at the same pitchas the word lines, and thus they are generally laid out substantially infour separate stages. In FIG. 19, each stage of the layout correspondsto the block (B₁ through B₄) of the word drivers. The layout area is notincreased if separate power lines are used for the blocks. Thus, thevalue of l can be made smaller than the memory cell count per data linepair. It will be clear that this value can be contrarily increased, andthus the degree of freedom of the value of l is large. Therefore, thesubthreshold current I_(A) upon operation can be minimized by properlysetting the l, (k·d) and (j·e).

The two-dimensional arrangement of word drivers has been mentionedabove. This arrangement can be applied to the following case.

FIG. 20 shows an example of the hierarchical type power line system ofthe same two-dimensional arrangement applied to decoders as in FIG. 16.An AND circuit is formed of two stages of a NAND circuit and inverter ofa CMOS logic circuit. The feature of this arrangement is that thehierarchical type power lines are provided on both sides of V_(cc) andV_(ss). All NAND circuits produce V_(cc) in the standby mode, and asmall number of the NAND circuits produce 0 V in the operating mode.Since the subthreshold current is determined by the n-channel MOStransistor on V_(ss) side, the hierarchical type power lines are used onV_(ss) side. On the contrary, all the inverters produce 0 V in thestandby mode, and a small number of ones of the inverters produce V_(cc)in the operating mode. Since the subthreshold current is determined bythe p-channel MOS transistor, the hierarchical type power lines are usedon V_(cc) side. Thus, since the hierarchical type power lines are usedon both sides of V_(cc) and V_(ss), use of multiple-stage logic circuitswill not make the operation unstable and the subthreshold current can bereduced.

Even in the circuit operating at around the central point of Vcc/2 suchas the sense amplifier drive circuit, this invention is applied to bothsides of V_(cc) and V_(ss), thus reducing the subthreshold current. Thisinvention can be applied to a circuit group if the circuits of thecircuit group output the same voltage in the standby mode, and if asmall number of ones of the circuits are operated in the operating mode.At this time, all the circuits are not necessary to be of the sametransistor size, but may be of different configurations. In addition,the number of circuits within the block and the number of blocks withinthe sector may be different.

When a plurality of circuits are simultaneously operated, a plurality ofcircuits within a single block may be operated or a plurality of blocksmay be selected at a time. In addition, the transistors operating asswitches may be grouped into a plurality of blocks and arrangedseparately. In this case, the interconnection resistance can be reducedby using power lines of a short length, and thus the power lines of theselected block can be charged in a short time.

Another example will be mentioned in which both the one-dimensionalselection and the two-dimensional selection can be made in onesemiconductor integrated circuit. FIG. 21 schematically shows anapplication of the invention to the hierarchical type word linestructure. The hierarchical type word line structure is described in1993 IEEE International Solid-State Circuits Conference, Digest ofTechnical Papers, pp. 50-51 (February 1993) or in Eighteenth EuropeanSolid State Circuits Conference, proceedings, pp. 131-134 (September1992).

The main drivers are formed of j blocks MB₁ through MB_(j) of one drivereach. The NMOS transistors Q_(M1) through Q_(Mj) are block selectiontransistors. The subword drivers are formed of J·k blocks of one drivereach in a matrix shape. In addition, these blocks are divided into jsectors SS₁ through SS_(j) of k blocks SB₁ through SB_(k) each. The PMOStransistors of row address drivers RLD₁ through RLD_(j) serve as sectorselecting transistors, and the PMOS transistors of subcolumn driversSCLD₁ through SCLD_(k) serve as block selecting transistors.

The operation will be mentioned. For example, the subcolumn driver SCLD₁is selected, and the subcolumn address line SCL₁ is driven, by the maincolumn address line MCL₁ and the row address line RL₁ driven by the rowaddress driver RLD₁. In addition, a subword driver is selected to driveone subword line SWL, and the memory cell MC is selected, by thesubcolumn address line SCL₁ and main word line MWL.

The subthreshold current after the subword line SWL is driven is, asdescribed above, reduced by the one-dimensional selection of the mainword driver block and the two-dimensional selection of the subworddriver block. In other words, this current is the sum of the current ofl·i_(M) flowing in the nonselected block MB₁ of the main word driver,the current of (j-1)·a·i_(M) flowing in the nonselected blocks MB₂through MB_(j) of the main word driver, the current of l·i_(S) flowingin the selected block SB₁ of the selected sector SS₁ of the subworddriver, the current of (k-1)·a·i_(S) flowing in the nonselected blocksSB₂ through SB_(k) of the selected sector SS₁ of the subword driver, andthe current of (j-1)·b·i_(S) flowing in the nonselected sectors SS₂through SS_(j) of the main word driver.

The reason why the block selection transistor of the main word driver isprovided as an NMOS transistor on the ground side is that since thenonselected main word driver produces V_(CH), the subthreshold currentis determined by the NMOS transistor on the ground side.

The subword drivers are originally two-dimensionally arranged, and theselection lines therefor serve both as themselves and as power lines.Thus, the two-dimensional selection of the invention can be easilyapplied to the subword drivers. Moreover, since the number is large, theeffect of reducing the subthreshold current is great. In this case, ifthe main word drivers are divided likewise, the one-dimensionalselection of the invention can be realized. In that case, since thesignal for selecting the blocks of the main word drivers is thecomplementary signal of the signal for selecting the sector of thesubword drivers, it can be generated by a common circuit.

The hierarchical type word line arrangement has the effect that thecharging and discharging currents can be reduced by decreasing thenumber of the memory cells to be selected as compared with theconventional word line construction. In addition, the hierarchical typeword line arrangement is suitable in the application of the selectivepower supplying of the invention for reducing the subthreshold current.

In this embodiment, the column address lines are hierarchicallyarranged. Therefore, the interconnection resistance and capacitance ofthe subcolumn address lines are small, and thus the operation speed canbe increased. One of the advantages of the hierarchical type word linestructure is that the interconnection delay of the subword lines issmall enough for high operation speed. In this embodiment, thehierarchical type word line structure is further improved for high speedoperation.

This invention can be applied not only to the DRAM, but also to thestatic random access memory (SRAM), the read only memory (ROM), theflush memory and the memory-incorporated logic LSI. This invention isalso applied to other logic circuits than the CMOS circuits such asn-channel MOS logic circuits. While a positive voltage of 2.5 V relativeto ground potential is applied as the operating voltage as above, anegative voltage relative to ground potential may be applied as theoperating voltage. This invention can be applied to a semiconductorintegrated circuit which is operated under a negative voltage of whichthe absolute value is lower than 2.5 V, or for example, under -2 V.

According to this invention, as will be obvious from the abovedescription, a semiconductor integrated circuit can be realized whichcan reduce the subthreshold current without decreasing the operatingspeed, and make high speed operation with a low consumption power. Theeffect of the invention is the greater, the less the threshold voltage.In the LSI in which the constant-current threshold voltage at which thesubthreshold current becomes dominant in the active current is less thanabout 0.2 V (the extrapolated threshold voltage is lower than about 0.4V), the effect is great. In other words, the threshold voltage isnecessary to be determined from the view point of the operating speedwhen the operating voltage is about 2.5 V or below or from the scalingrule when the gate length is about 0.25 μm or below, and hence theeffect is extremely large in that LSI.

What is claimed is:
 1. A method for supplying voltages to a semiconductor device which operates at operating voltages of a first voltage and a second voltage and which includes a semiconductor logic circuit including a MOS transistor having a source-drain path disposed between a first electric source line and a second electric source line, wherein a leak current flows between the source and drain of said MOS transistor even on a condition that a gate voltage is equal to a source voltage, comprising the steps of:supplying said second voltage to said second electric source line; supplying said first voltage to said first electric source line when said semiconductor logic circuit is in an active condition; and supplying said second voltage to said first electric source line when said semiconductor logic circuit is in a non-active condition.
 2. A method for supplying voltages to a semiconductor device according to claim 1,wherein a source of said MOS transistor is connected to said first electric source line.
 3. A method for supplying voltages to a semiconductor device according to claim 1,wherein a voltage difference between said first voltage and said second voltage is equal to or less than 2.5 V.
 4. A method for supplying voltages to a semiconductor device according to claim 1,wherein a threshold voltage of said MOS transistor is equal to or less than 0.2 V when said threshold voltage is defined as a voltage between a gate and said source caused by a drain current of a constant absolute value of 10 nA at a room temperature on a condition that a ratio of an effective gate width to an effective gate length is
 30. 5. A method for supplying voltages to a semiconductor device according to claim 1,wherein a thickness of a gate oxide film of said MOS transistor is equal to or less than 6 nm.
 6. A method for supplying voltages to a semiconductor device according to claim 1,wherein an effective channel length of said MOS transistor is equal to or less than 0.25 μm.
 7. A method for supplying voltages to a semiconductor device according to claim 1,wherein said semiconductor logic circuit is comprised of CMOS logic gates.
 8. A semiconductor integrated circuit device comprising:a first electric source line supplied with a first voltage; a second electric source line supplied with a second voltage; a circuit block including:a first electric source node; a second electric source node connected to said second electric source line; and a MOS transistor in which a leak current flows between a drain and a source even in a condition that a gate voltage is equal to a source voltage; a first switching element provided between said first electric source line and said first electric source node of said circuit block; and a second switching element provided between said second source line and a said first electric source node of said circuit block.
 9. A semiconductor integrated circuit device according to claim 8,wherein said source of said MOS transistor is connected to said first electric source line.
 10. A semiconductor integrated circuit device according to claim 8,wherein a voltage difference between said first voltage and said second voltage is equal to or less than 2.5 V.
 11. A semiconductor integrated circuit device according to claim 8,wherein a threshold voltage of said MOS transistor is equal to or less than 0.2 V when said threshold voltage is defined as a voltage between a gate and said source caused by a drain current of a constant absolute value of 10 nA at a room temperature on a condition that a ratio of an effective gate width to an effective gate length is
 30. 12. A semiconductor integrated circuit device according to claim 8,wherein a thickness of a gate oxide film of said MOS transistor is equal to or less than 6 nm.
 13. A semiconductor integrated circuit device according to claim 8,wherein an effective channel length of said MOS transistor is less than 0.25 μm.
 14. A semiconductor integrated circuit device according to claim 8,wherein said circuit block is comprised of CMOS logic gates.
 15. A semiconductor integrated circuit device according to claim 8,wherein said first switching element includes a MOS transistor.
 16. A semiconductor integrated circuit device according to claim 15,wherein an absolute value of said threshold voltage of said MOS transistor of said first switching element is larger than that of said MOS transistor of said circuit block, when said threshold voltage is defined as a voltage between a gate and said source caused by a drain current of a constant absolute value of 10 nA at a room temperature on condition that a ratio of an effective gate width to an effective gate length is
 30. 17. A semiconductor integrated circuit device according to claim 15,wherein a polarity of said MOS transistor of said first switching element is the same as that of said MOS transistor of said circuit block.
 18. A semiconductor integrated circuit device according to claim 15,wherein said second switching element includes a MOS transistor.
 19. A semiconductor integrated circuit device according to claim 15,wherein a polarity of said MOS transistor of said first switching element is complementary to that of said MOS transistor of said second switching element. 